Low-voltage push-pull voltage converters. Hello student

The most widespread are push-pull secondary power supplies, although they have a more complex electrical circuit than single-ended ones. They make it possible to obtain significantly higher output power with high efficiency.
Circuits of push-pull converters-inverters have three types of switching on key transistors and the primary winding of the output transformer: half-bridge, bridge and with a primary winding having a tap from the middle.

Half-bridge scheme for constructing a key cascade.
Its feature is the inclusion of the primary winding of the output transformer at the midpoint of the capacitive divider C1 - C2.

The amplitude of the voltage pulses at the junctions of the emitter-collector transistors T1 and T2 does not exceed Upit of the supply voltage. This allows the use of transistors with a maximum voltage Ueq up to 400 volts.
At the same time, the voltage on the primary winding of the transformer T2 does not exceed the value Upit / 2, because it is removed from the divider C1 - C2 (Upit / 2).
The control voltage of opposite polarity is supplied to the bases of the key transistors T1 and T2 through the transformer Tr1.


IN pavement converter capacitive divider (C1 and C2) is replaced by transistors T3 and T4. The transistors in each half-cycle open in pairs along the diagonal (T1, T4) and (T2, T3).

The voltage at the junctions Ueq of closed transistors does not exceed the supply voltage Upit. But the voltage on the primary winding of the transformer Tr3 will increase and will be equal to the value of Upit, which increases the efficiency of the converter. The current through the primary winding of the transformer Tr3 at the same power, in comparison with the half-bridge circuit, will be less.
Due to the difficulty in setting up control circuits of transistors T1 - T4, the bridge switching circuit is rarely used.

Inverter circuit with the so-called push-pull output is most preferable in powerful converters-inverters. A distinctive feature in this circuit is that the primary winding of the output transformer Tr2 has an output from the middle. For each half-cycle of the voltage, one transistor and one half-winding of the transformer alternately operate.

This circuit is characterized by the highest efficiency, low ripple and low noise emission. This is achieved by reducing the current in the primary winding and reducing the dissipated power in the key transistors.
The amplitude of the voltage pulses in the half of the primary winding Tr2 increases to the value of Upit, and the voltage Ueq on each transistor reaches the value of 2 Upit (EMF of self-induction + Upit).
It is necessary to use transistors with a high value of Ukemax, equal to 600 - 700 volts.
The average current through each transistor is equal to half the current consumption from the supply network.

Current or voltage feedback.

A feature of push-pull circuits with self-excitation is the presence feedback (OS) from output to input, current or voltage.

In the scheme current feedback the coupling winding w3 of the transformer Tr1 is connected in series with the primary winding w1 of the output transformer Tr2. The greater the load at the output of the inverter, the greater the current in the primary winding Tr2, the greater the feedback and the greater the base current of the transistors T1 and T2.
If the load is less than the minimum permissible, the feedback current in the winding w3 of the transformer Tr1 is insufficient to control the transistors and the generation of alternating voltage is interrupted.
In other words, when the load is lost, the generator does not work.

In the scheme voltage feedback the feedback winding w3 of the transformer Tr2 is connected through a resistor R with the coupling winding w3 of the transformer Tr1. This circuit provides feedback from the output transformer to the input of the control transformer Tr1 and then to the base circuits of transistors T1 and T2.
Voltage feedback is weakly load dependent. If there is a very large load at the output (short circuit), the voltage on the winding w3 of the transformer Tr2 decreases and a moment may come when the voltage on the base windings w1 and w2 of the transformer Tr1 will be insufficient to control the transistors. The generator will stop working.
Under certain circumstances this phenomenon can be used as output short-circuit protection.
In practice, both circuits with feedback of the OS both in current and in voltage are widely used.

Push-pull inverter circuit with voltage feedback

For example, consider the operation of the most common converter-inverter circuit - a half-bridge circuit.
The scheme consists of several independent blocks:

• - a rectifier unit - converts an alternating voltage of 220 volts 50 Hz into a constant voltage of 310 volts;
• - device for triggering pulses - generates short voltage pulses to start the generator;
• - AC voltage generator - converts a constant voltage of 310 volts into an alternating voltage of a rectangular shape high frequency 20 - 100 kHz;
• - rectifier - converts alternating voltage 20 -100 kHz into direct voltage.

Immediately after turning on the 220 volt power supply, the triggering pulse device begins to work, which is a sawtooth voltage generator (R2, C2, D7). From it, the triggering pulses are fed to the base of the transistor T2. The autogenerator starts up.
Key transistors open in turn and in the primary winding of the output transformer Tr2, included in the diagonal of the bridge (T1, T2 - C3, C4), a rectangular AC voltage is formed.
The output voltage is removed from the secondary winding of the transformer Tr2, rectified by diodes D9 - D12 (full-wave rectification) and smoothed by capacitor C5.
The output is a constant voltage of a given value.
Transformer T1 is used to transmit feedback pulses from the output transformer Tr2 to the bases of key transistors T1 and T2.


A push-pull UPS circuit has several advantages over a single-ended circuit:

• - the ferrite core of the Tr2 output transformer works with active magnetization reversal (the magnetic core is most fully used in terms of power);
• - collector - emitter voltage Ueq on each transistor does not exceed the voltage of the direct current source of 310 volts;
• - when the load current changes from I \u003d 0 to Imax, the output voltage changes slightly;
• - high voltage surges in the primary winding of the transformer Tr2 are very small, respectively, the level of radiated interference is less.

And one more remark in favor of the push-pull circuit !!

Let's compare the work of a two-stroke and one-stroke autogenerators with the same load.
Each key transistor T1 and T2 for one cycle of the generator is used only half the time (one half-wave), the second half of the cycle "rests". That is, all the generated power of the generator is divided in half between both transistors and the transfer of energy to the load goes continuously (either from one transistor, then from another), during the entire cycle. Transistors work in a sparing mode.
In a single-cycle generator, the accumulation of energy in the ferrite core occurs during half the cycle, in the second half of the cycle it is returned to the load.

A key transistor in a single-ended circuit operates four times more intensely than a key transistor in a push-pull circuit.

Perhaps one of the simplest voltage converter circuits is a simple push-pull converter on field-effect transistors, which are connected according to the multivibrator circuit. Zener diodes can be excluded from the circuit, unless, of course, the circuit is designed to be powered from a voltage of no more than 12 volts. The resistors in the circuit are not critical, their rating can be in the regions from 220 ohms to 1 kilo-ohm, they limit the gate current of the field-effect transistors, therefore, by selecting their rating, you can adjust the frequency of the converter. It is desirable to use resistors with a power of 0.5-1 watts, overheating of these resistors is possible, but this is not scary.

The operation of a push-pull converter is quite simple, the transistors alternately opening and closing create an alternating high-frequency voltage in the primary winding of the transformer. The transformer is wound on a yellow ferrite ring made of computer unit supply, although 2000HM rings can be used.

To power the LDS, the transformer in the primary winding contains 6 turns with a tap from the middle, the wire is 0.6-1 mm, the secondary winding contains 90 turns and is stretched along the entire ring, the wire is 0.2-0.4 mm, insulation can be omitted if for the primary device, use a stranded wire in rubber insulation.

The converter is capable of developing a power of up to 20 watts when using field-effect transistors of the IRFЗ44 series and up to 30 watts if using transistors of the IRF3205 type. The scope of application of this kind of push-pull converters is quite wide, since the converter is capable of developing a good output power and has very compact dimensions, it is advisable to use it for charging capacitors or for powering an LDS in field conditions, where there is no 220 volt household network, to supply active devices with such a converter - receivers, low-power chargers are not allowed, since the frequency of the converter is high enough.

In fact, the tip of the soldering iron is hardened due to a short circuit. The secondary winding contains half a turn, the voltage of the strand is 1 volt, but the current strength reaches 15 Amperes! It is because of the reduced voltage that the load is not so great, during operation the parts are almost cold.

Schematic diagrams of simple voltage converters based on autogenerators, built using transistors.

Self-excited generators (self-excited generators) usually use positive feedback to excite electrical oscillations. There are also autogenerators based on active elements with negative dynamic resistance, but they are practically not used as converters.

Single stage voltage converters

Most simple circuit a single-stage voltage converter based on an oscillator is shown in Fig. 1. This type of generators is called blocking generators. The phase shift to ensure the conditions for the occurrence of oscillations in it is provided by a certain turn on of the windings.

Figure: 1. Circuit of a voltage converter with transformer feedback.

An analogue of the 2N3055 transistor - KT819GM. The blocking generator allows you to obtain short pulses with a high duty cycle. In shape, these impulses approach rectangular.

The capacities of the oscillatory circuits of the blocking generator are, as a rule, small and are due to interturn capacities and the installation capacitance. The limiting frequency of the blocking generator generation is hundreds of kHz. The disadvantage of this type of generator is the pronounced dependence of the generation frequency on changes in the supply voltage.

The resistive divider in the base circuit of the transistor of the converter (Fig. 1) is designed to create an initial bias. A somewhat modified version of the converter with transformer feedback is shown in Fig. 2.

Figure: 2. Diagram of the main (intermediate) block of a high-voltage source based on an autogenerating converter.

The oscillator operates at a frequency of approximately 30 kHz. At the output of the converter, a voltage is formed with an amplitude of up to 1 kV (determined by the number of turns of the step-up winding of the transformer).

Transformer T1 is made on a dielectric frame inserted into the B26 armor core made of M2000NM1 (M1500NM1) ferrite. The primary winding contains 6 turns; secondary winding - 20 turns of PELSHO wire with a diameter of 0.18 mm (0.12 ... 0.23 mm).

The step-up winding to achieve an output voltage of 700 ... 800 V has approximately 1800 turns of PEL wire with a diameter of 0.1 mm. Every 400 turns during winding, a dielectric spacer made of capacitor paper is laid, the layers are impregnated with capacitor or transformer oil. The coil terminals are filled with paraffin.

This converter can be used as an intermediate one for supplying subsequent stages of high voltage generation (for example, with electric spark gaps or thyristors).

The next voltage converter (USA) is also made on one transistor (Fig. 3). The base bias voltage is stabilized by three series connected diodes VD1 - VD3 (forward bias).

Figure: 3. Diagram of a voltage converter with transformer feedback.

The collector junction of the transistor VT1 is protected by a capacitor C2, in addition, a chain of diode VD4 and Zener diode VD5 is connected in parallel to the collector winding of transformer T1.

The generator produces pulses close to rectangular in shape. The generation frequency is 10 kHz and is determined by the value of the capacitor C3. An analogue of the 2N3700 transistor - KT630A.

Push-pull voltage converters

The diagram of a push-pull transformer voltage converter is shown in Fig. 4. Analog of the transistor 2N3055 - KT819GM. The transformer of a high-voltage converter (Fig. 4) can be made using an open-ended ferrite core of a round or rectangular cross-section, as well as on the basis of a television line transformer.

When using a round ferrite core with a diameter of 8 mm, the number of turns of the high-voltage winding, depending on the required output voltage, can reach 8000 turns of wire with a diameter of 0.15 ... 0.25 mm. The collector windings contain 14 turns of wire with a diameter of 0.5 ... 0.8 mm.

Figure: 4. Diagram of a push-pull converter with transformer feedback.

Figure: 5. A variant of the high-voltage converter circuit with transformer feedback.

The feedback windings (base windings) contain 6 turns of the same wire. When connecting the windings, their phasing should be observed. The output voltage of the converter is up to 8 kV.

Transistors of domestic production, for example, KT819 and the like, can be used as transistors of the converter.

A variant of the circuit of a similar voltage converter is shown in Fig. 5. The main difference lies in the bias supply circuits to the transistor bases.

The number of turns of the primary (collector) winding is 2x5 turns with a diameter of 1.29 mm, of the secondary - 2x2 turns with a diameter of 0.64 mm. The output voltage of the converter is entirely determined by the number of turns of the step-up winding and can reach 10 ... 30 kV.

A. Chaplygin's voltage converter does not contain resistors (Fig. 6). It is powered by a 5-6 battery and is capable of delivering up to 1 A at a voltage of 12 V.

Figure: 6. Schematic of a simple, high-efficiency voltage converter powered by a 5V battery.

The rectifier diodes are transistor transitions of the oscillator. The device is also capable of operating at a supply voltage reduced to 1 V.

For low-power converter options, you can use transistors such as KT208, KT209, KT501 and others. The maximum load current should not exceed the maximum base current of the transistors.

Diodes VD1 and VD2 are optional, but they allow you to get an additional voltage of 4.2 V at the output of negative polarity. The efficiency of the device is about 85%. The T1 transformer is made on the K18x8x5 2000NM1 ring. Windings I and II have 6 each, III and IV - 10 turns of wire PEL-2 0.5.

Inductive three-point converter

The voltage converter (Fig. 7) is made according to the inductive three-point scheme and is intended for measuring high-resistance resistance and allows you to obtain an unstabilized voltage of 120 ... 150 V at the output.

The current consumed by the converter is about 3 ... 5 mA at a supply voltage of 4.5 V. The transformer for this device can be created on the basis of the BTK-70 television transformer.

Figure: 7. Voltage converter circuit according to the inductive three-ton circuit.

Its secondary winding is removed, instead of it, a low-voltage winding of the converter is wound - 90 turns (two layers of 45 turns each) of the PEV-1 wire 0.19 ... 0.23 mm. Branch from the 70th turn from the bottom according to the scheme. Resistor R1 - 12 ... 51 kOhm.

Voltage converter 1.5 V / -9 V

Figure: 8. Voltage converter circuit 1.5 V / -9 V.

The converter (Fig. 8) is a single-cycle relaxation generator with capacitive positive feedback (C2, C3). A step-up autotransformer T1 is included in the collector circuit of the VT2 transistor.

The converter uses the reverse connection of the rectifier diode VD1, i.e. when the transistor VT2 is open, the supply voltage Un is applied to the winding of the autotransformer, and a voltage pulse appears at the output of the autotransformer. However, the diode VD1, turned on in the opposite direction, is closed at this time, and the load is disconnected from the converter.

At the moment of pause, when the transistor closes, the polarity of the voltage across the windings of T1 is reversed, the diode VD1 opens, and the rectified voltage is applied to the load.

In subsequent cycles, when the transistor VT2 is locked, the filter capacitors (C4, C5) are discharged through the load, providing a direct current flow. The inductance of the step-up winding of the autotransformer T1 in this case plays the role of a smoothing filter choke.

To eliminate the magnetization of the autotransformer core by direct current of the transistor VT2, the magnetization of the autotransformer core is used due to the inclusion of capacitors C2 and C3 in parallel to its winding, which are simultaneously a feedback voltage divider.

When the transistor VT2 closes, the capacitors C2 and C3 during the pause are discharged through a part of the transformer winding, remagnetizing the core T1 by the discharge current.

The generation frequency depends on the voltage at the base of the VT1 transistor. The stabilization of the output voltage is carried out by means of negative feedback (OOS) for constant voltage by means of R2.

With a decrease in the output voltage, the frequency of the generated pulses increases with approximately the same duration. As a result, the frequency of recharging the filter capacitors C4 and C5 increases and the voltage drop across the load is compensated. With an increase in the output voltage, the generation frequency, on the contrary, decreases.

So, after charging the storage capacitor C5, the generation frequency drops dozens of times. Only rare pulses remain, which compensate for the discharge of the capacitors at rest. This stabilization method made it possible to reduce the quiescent current of the converter to 0.5 mA.

Transistors T1 and иметьT2 should have as high a gain as possible to improve efficiency. The autotransformer winding is wound on a K10x6x2 ferrite ring made of 2000NM material and has 300 turns of PEL-0.08 wire with a tap from the 50th turn (counting from the "grounded" terminal). The VD1 diode must be high frequency and have a low reverse current. The adjustment of the converter is reduced to setting the output voltage equal to -9 V by selecting the resistor R2.

Voltage converter with PWM control

In fig. 9 shows a diagram of a stabilized voltage converter with pulse width control. The converter remains operational when the battery voltage decreases from 9 ... 12 to 3V. Such a converter turns out to be most suitable for battery-powered equipment.

The efficiency of the stabilizer is at least 70%. Stabilization is maintained when the power supply voltage drops below the output stabilized voltage of the converter, which cannot be provided by a traditional voltage regulator. The stabilization principle used in this voltage converter.

Figure: 9. Scheme of a stabilized voltage converter with PWM control.

When the converter is turned on, the current through the resistor R1 opens the transistor ѴT1, the collector current of which, flowing through the winding II of the transformer T1, opens powerful transistor ѴT2. Transistor ѴT2 enters the saturation mode, and the current through the winding I of the transformer increases linearly.

Energy is stored in the transformer. After some time, the transistor ѴT2 goes into active mode, self-induction EMF arises in the transformer windings, the polarity of which is opposite to the voltage applied to them (the transformer magnetic circuit is not saturated).

The transistor ѴT2 closes like an avalanche and the self-induction EMF of the winding I through the diode VD2 charges the capacitor C3. Capacitor C2 contributes to a clearer closing of the transistor. Then the process is repeated.

After a while, the voltage across the capacitor СЗ increases so much that the Zener diode VD1 opens, and the base current of the transistor ѴТ1 decreases, while the base current decreases, and hence the collector current of the transistor Т2.

Since the energy accumulated in the transformer is determined by the collector current of the transistor ѴT2, a further increase in the voltage across the capacitor C3 stops. The capacitor is discharged through the load. Thus, a constant voltage is maintained at the output of the converter. The output voltage is set by the Zener diode VD1. The conversion frequency varies within 20 ... 140 kHz.

Voltage converter 3-12V / + 15V, -15V

A voltage converter, the circuit of which is shown in Fig. 10 differs in that the load circuit is galvanically isolated from the control circuit. This makes it possible to obtain several secondary stable voltages. The use of an integrating link in the feedback circuit improves the stabilization of the secondary voltage.

Figure: 10. Scheme of a stabilized voltage converter with a bipolar output 15 + 15V.

Conversion frequency decreases almost linearly with decreasing supply voltage. This circumstance enhances the feedback in the converter and increases the stability of the secondary voltage.

The voltage across the smoothing capacitors of the secondary circuits depends on the energy of the pulses received from the transformer. The presence of resistor R2 makes the voltage across the storage capacitor C3 dependent on the pulse repetition rate, and the degree of dependence (steepness) is determined by the resistance of this resistor.

Thus, the trimming resistor R2 can set the desired dependence of the change in the voltage of the secondary windings on the change in the supply voltage. Field-effect transistor ѴT2 - current stabilizer. The efficiency of the converter can be up to 70 ... 90%.

The instability of the output voltage at a supply voltage of 4 ... 12 V is not more than 0.5%, and when the ambient temperature changes from -40 to + 50 ° C - not more than 1.5%. The maximum load power is 2 W.

When adjusting the converter, the resistors R1 and R2 are set to the position of the minimum resistance and connect the equivalent loads RH. A supply voltage of 12 V is supplied to the device input, and with the help of a resistor R1, a voltage of 15 V is set at the load Rn. Further, the supply voltage is reduced to 4V and the output voltage is also 15 V with resistor R2. Repeating this process several times, a stable output voltage is achieved.

Windings I and II and the magnetic circuit of the transformer are the same for both converters. The windings are wound on a B26 armored magnetic core made of 1500NM ferrite. Winding I contains 8 turns of PEL 0.8 wire, and II - 6 turns of PEL 0.33 wire (each of windings III and IV consists of 15 turns of 0.33 mm PEL wire).

Small-sized mains voltage converter

A diagram of a simple small-sized mains voltage converter made of available elements is shown in Fig. 11. The device is based on a conventional blocking generator on a transistor VT1 (KT604, KT605A, KT940).

Figure: 11. The circuit of the step-down voltage converter based on the blocking generator.

Transformer T1 is wound on an armored core B22 made of M2000NN ferrite. Windings Ia and Ib contain 150 + 120 turns of 0.1 mm PELSHO wire. Winding II has 40 turns of 0.27 mm PEL wire III - 11 turns of 0.1 mm PELSHO wire. First, winding Ia is wound, then - II, after - winding lb, and, finally, winding III.

The power supply is not afraid of a short circuit or an open circuit in the load, but it has a large voltage ripple factor, low efficiency, low output power (up to 1 W) and a significant level of electromagnetic interference. You can also power the converter from a direct current source with a voltage of 120 6. In this case, the resistors R1 and R2 (as well as the diode VD1) should be excluded from the circuit.

Low-current voltage converter for 440V

A low-current voltage converter for powering a Geiger-Muller gas-discharge counter can be assembled according to the diagram in Fig. 12. The converter is a transistor blocking generator with an additional step-up winding. Pulses from this winding charge the capacitor C3 through the rectifier diodes VD2, VD3 to a voltage of 440 V.

Capacitor SZ must be either mica or ceramic, for an operating voltage of at least 500 V. The duration of the blocking generator pulses is approximately 10 μs. The pulse repetition rate (tens of Hz) depends on the time constant of the circuit R1, C2.

Figure: 12. Diagram of a low-current voltage converter for powering a Geiger-Muller gas-discharge counter.

The magnetic core of the T1 transformer is made of two K16x10x4.5 3000NM ferrite rings glued together and insulated with a layer of varnish, Teflon or fluoroplastic.

At the beginning, winding III is wound in bulk - 420 turns of wire PEV-2 0.07, filling the magnetic circuit evenly. A layer of insulation is applied over the winding III. Windings I (8 turns) and II (3 turns) are wound with any wire over this layer, they should also be distributed as evenly as possible around the ring.

Pay attention to the correct phasing of the windings, it must be done before the first start. With a load resistance of the order of units of MΩ, the converter consumes a current of 0.4 ... 1.0 mA.

Voltage converter for powering the flash

The voltage converter (Fig. 13) is designed to power the flash unit. The transformer T1 is made on a magnetic circuit of two K40x28x6 permalloy rings folded together. The winding of the collector circuit of the transistor VT1 has 16 turns of PEV-2 0.6 mm; its base circuit is 12 turns of the same wire. The step-up winding contains 400 turns of PEV-2 0.2.

Figure: 13. Diagram of the voltage converter for the flash.

HL1 neon lamp is used from lamp starter daylight... The output voltage of the converter smoothly rises on the flash capacitor to 200 V in 50 seconds. In this case, the device consumes current up to 0.6 A.

Voltage converter PN-70

The PN-70 voltage converter, which is the basis of the device described below, is intended to power the flash lamps (Fig. 14). Typically, inverter battery power is consumed with minimum efficiency.

Regardless of the frequency of the flashes of light, the generator works continuously, consuming a large amount of energy and discharging the batteries.

Figure: 14. Diagram of the modified voltage converter PN-70.

O. Panchik succeeded in transferring the operation of the converter to the standby mode, who turned on the resistive divider R5, R6 at the output of the converter and sent a signal from it through the Zener diode VD1 to an electronic switch made on transistors VT1 - ѴТЗ according to the Darlington scheme.

As soon as the voltage across the flash capacitor (not shown in the diagram) reaches the nominal value determined by the value of the resistor R6, the Zener diode VD1 will break through, and the transistor switch will disconnect the 9 V battery from the converter.

When the voltage at the output of the converter decreases as a result of self-discharge or discharge of the capacitor to the flash lamp, the Zener diode VD1 will stop conducting current, the switch will turn on and, accordingly, the converter. Transistor ѴT1 must be installed on a copper heatsink with dimensions 50x22x0.5 mm.

What does an embedder do when he has nothing to do? Examines push-pull autogenerating converters, of course! To do, in fact, there is something, and a lot, but something lazy. Therefore, today I will still investigate the push-pull autogenerating converter. Like this: As in the picture above, they are drawn in books, but I don't like this drawing; not only does the converter look like a multivibrator in this style (which is far from the true principle of its operation), but also the output is on top (in the first picture, I slightly corrected it). Therefore, I offer my own version:
The picture runs a little ahead - where all these numbers came from, I will explain in the course of the article. First, let's look at the general principle of the circuit. When power is applied, the first to open is that transistor, the base-emitter voltage of which is less or the current transfer coefficient of which is greater (there are no completely identical transistors in nature). Let it be T2. Then a rising current will begin to flow through winding B. In this case, windings A and B together work as an autotransformer, as a result of which a voltage will be applied to the base T2 through the resistor R2, even higher than the supply voltage. This guarantees saturation of the transistor (since both junctions, collector and emitter, are open). In this case, T1 is closed, because the voltage at the collector of saturated T2 is low. T2 is open, the current through winding B is increasing, everything is great. However, this will continue exactly until the magnetic core of the transformer enters saturation. As soon as this happens, the inductances of the windings will drop sharply, and, therefore, the current through them will begin to tend to infinity, limited practically only by the resistance of the winding. Indeed, after all

UPD: In more detail and correctly, I have analyzed the operation of this scheme.

Like everything on earth, such a converter has its pros and cons. The first and most obvious plus is its fantastic simplicity. Only four parts are required, not including the transformer. Also, the pluses include the fact that the transformer in such a converter will never saturate too far, which limits losses. In addition, this is a real push-pull circuit, so the transformer does not need a gap, which means that rings from savings banks can be used, for example (which I am going to do next). With all the pluses, this scheme also has enough minuses. First, the magnetic circuit will still enter saturation, so there will be losses that could have been avoided. Secondly, it can be seen that the possibility of operation of such a converter is closely tied to the real properties of the magnetic circuit of the transformer (the error of indicating which in datasheets reaches 30%) and a little to the imperfection of transistors. I.e, calculate such a converter is impossible - its parameters can only be roughly estimated, well, or measured on a real circuit. The operating frequency will be determined by how quickly the magnetic circuit will enter saturation, that is, it will depend on the input voltage. Above I talked about rings from savings. For a toroidal core, the expression for the induction in the magnetic circuit is as follows: where μ is the magnetic permeability of the ring, μ 0 is the magnetic constant, N is the number of winding turns, I is the current in the winding, R is the radius of the ring. The rate of rise of the current in the winding is proportional to the applied voltage (see the very first formula), that is, the rate of rise of the magnetic flux will also be proportional to it, that is, the operating frequency will depend on the input voltage. In this case, the absolute value of the induction will be proportional to the product of the number of turns by the current, therefore the no-load current will be determined by the number of turns in windings A and B (the more turns, the less current saturation will be achieved). Hence, another drawback follows - in order to obtain a small no-load current, it is necessary to wind a lot of wire, which is especially tiring in the case of a toroidal core. Well, the no-load current will also depend on the applied voltage. From all that has been said, we can conclude that such a scheme is suitable when the simplicity of the converter outweighs the need for accurate predictability and quality of its characteristics. For example, in the case when the goal is to have a little fun on a spring evening.

Let's move from theory to practice. In my bins was an unidentified ring taken from a savings bank. Its diameter is 10 mm, height - 3.5 mm, thickness - 2 mm. That is, it looks like an EPCOS R 10 x 6 x 4 ring.
I wound 10 turns of wire around it and measured the inductance of the resulting coil. It turned out 286 μH, which corresponds to a permeability of about 8000. That is, according to the datasheet above, the material of the ring is either T37 or T38. Their saturation induction is something around 400 mT. I figured that I would not be too lazy to wind no more than 15 turns. According to the second formula, you can calculate that the saturation current will be something around 65 mA. Fine; fits well into the capabilities of the main "just transistors" - BC547 / 847/817. After that, I wound the windings - primary, 15 turns in two wires, and secondary, 63 turns (how many I mastered). The transformation ratio is 4.2, that is, from 1.5 V we get about 6.3 V.
Assembled a diagram. I put 510 Ohm resistors in the base of the transistors (which I found). At the same time, with a minimum input voltage (I took a minimum of 0.9 V with an eye on the battery as a source), the base current will be sufficient to provide a collector current sufficient to saturate the transformer with a minimum current transfer ratio of transistors (traditionally 100). 65 mA). Collected:
Submitted 1.5 V. It worked!
The output is 6.3 V RMS, exactly as designed. You can put a rectification circuit with a doubling and get 12 V. The voltage on the collectors:
It can be seen that the amplitude of the pulses is 3 V, that is, twice the supply voltage. So practice really coincides with theory - the primary winding works like an autotransformer. Base voltages (do not trust the frequency measurement, the oscilloscope glitches due to overshoots; the time grid is the same as above):
The consumed current. I measured the voltage across a 10 Ohm resistor connected in series with the converter:
About 76 mA peak. According to the second formula, the saturation induction can be calculated - it turns out about 457 mT, that is, ferrite, apparently, is still T38. The average no-load current at a voltage of 1.5 V was about 30 mA. The converter starts at an input voltage of 0.5 V. As for me, such a circuit is a great way to use rings from savings in simple converters 1.5 - 5 V / 3.3 V. Of course, it would be nice to put a stabilizer at the output (with a diode bridge, of course), in the simplest case is linear, the same L78L33. The efficiency of such a solution will not be particularly high, but in terms of cost and simplicity it will probably bypass even Chinese products.

Timing charts

When choosing a scheme for constructing a switching power supply, the developer is primarily guided by the expected overall dimensions and simplicity of circuitry solutions. Network sources supplying loads of low power (up to 100-150 W), built-in into sufficiently large equipment, are best built using a single-cycle fly-back scheme. For stabilizers that do not require galvanic isolation of the load from the mains, a chopper circuit is used. When powered by galvanic cells or batteries, you can use a booster circuit. However, situations are not excluded in which the listed converters and stabilizers cannot be used.

Case one- the device, powered from the AC mains, has limited dimensions (for example, it is not possible to place a sufficiently large storage transformer in the converter flask in the device case).

Second case- - the power consumption of the device exceeds 150 ... 200W.

Third case- individual parts of the device circuit require additional power supply, galvanically isolated from the rest of the circuit.

In all these cases, the development of so-called two-strokecircuits of converters with galvanic isolation of the primary and secondary circuits. The most widespread among push-pull converters are three schemes: two-phase push-pull, half-bridge and full-bridge. The advantage of these circuits is that, if necessary, the developer can easily introduce an output voltage stabilization unit into the design, or refuse it. In the first case, the converter will be a full-fledged power source to which you can connect any load. In the second case, a simple converter of electrical energy will be obtained, requiring additional stabilization at the output. In some cases, such a simple converter will suit the developer. Since all three schemes of push-pull converters have many analogies, we will talk about them in one chapter, focusing on individual characteristics and conducting a comparative analysis.

Push-pull two-phase scheme

Figure: 14.1.Basic push-pull push-pull converter circuit

This circuit (Figure 14.1) consists of two key elements, which are powerful bipolar or field-effect transistors. The transformer Tr has a primary and secondary winding, divided into half windings. A power supply terminal is connected to the midpoint of the primary winding. The secondary circuit is a two-phase full-wave rectifier VD1, VD2, as well as a ripple filter (in this circuit, the filter element is a capacitor C f).

In the first measure, as shown in Fig. 14.2, l is closed, Kl2 is open, the current flows through the half-winding 1.1 and is transformed into a half-winding 2.1. Diode VD1 is open and conducts current i 2.1, recharging capacitor Сф. In the second measure shown in Fig. 14.3, the key Kl. L is closed and the key Kl2 is opened. Accordingly, the current i 1.2 flows through
semi-winding 1.2 and transformed into semi-winding 2.2. Diode VD1 is locked, diode VD2 conducts current i 2 2, recharging capacitor C f.

Thus, the transfer of energy to the load occurs during both cycles.

To move on to the parameters of real circuits, we first assume that we nevertheless have the possibility of using ideal elements. That is, the transistors can instantly switch, there is no reverse recovery time of the diodes, the primary winding has a very large value of the magnetizing inductance (according to the equivalent circuit). Under these conditions, it is very easy to determine the dependence of the output voltage on the input voltage. The voltage of the primary winding is transformed into a secondary winding without losses, with a transformation ratio:

Transformation ratios n land n 2are considered the same, moreover, they equalize the number of turns of the primary and secondary half-windings:

Voltage on the primary winding in the closed key mode (without taking into account the voltage drop across the power switch):

Since the circuit is built with full-wave rectification at the output, the relationship between supply voltage and voltage across the load is:

It is not yet clear to us how the voltage regulation on the load can be introduced. Therefore, it is necessary to remember the fill factor and extend it to a push-pull circuit. Let's try to figure out what happens if we narrow the control impulses, as shown in Fig. 14.4. The filling factor in the case of a push-pull circuit is determined in the same way as for a single-cycle one:

where γ is the ratio of the open state time of one key to the switching period.

Figure:14.4. To the determination of the fill factor

In this case, we determine the duty cycle for one arm of the push-pull circuit. ... Let us determine the average value of the load current, taking into account that the transfer of energy is carried out during both half-periods, which means that the average voltage value for one cycle of operation must be doubled:

Figure: 14.5.Graphs explaining the operation of the push-pull converter circuit

Thus, by adjusting γ in the range from 0 to 0.5, it is possible to linearly regulate the voltage across the load. In a real circuit, in no case should the converter be allowed to operate with γ \u003d 0.5. Typical γ value should not exceed 0.4 ... 0.45. The thing is that the elements used cannot have ideal properties. As we know, the primary winding has a limited inductance L μ, which stores energy:

The maximum current i μ shown in the graph (Fig.14.7) is determined from the ratio:

When CL1 is opened, the energy accumulated in the magnetic circuit tends to maintain the current. If the circuit did not have a protective diode VDp 2 shown in Fig. 14.6, a negative voltage surge would appear on Cl2. The ability of bipolar transistors to withstand negative voltage surges is small (units of volts), therefore, the discharge current i μ must be closed through the diode VDp 2. The diode practically "short-circuits" the winding ω 2 2 and quickly discharges L μ (Fig. 14.8). During the discharge, thermal energy is released, which can be taken into account through the following ratio:

Figure:14.6. To an explanation of the switching

processes in a real push-pull scheme

converter Figure: 14.7.Determination of the magnetizing current

Figure: 14.8.Discharge inductance magnetization

When the push-pull converter is operating, the discharge diodes are switched on alternately. It should also be remembered that these diodes already exist as part of MOSFET transistors, as well as some IGBT transistors, so there is no need to introduce additional elements.

The second trouble is associated with the final recovery time of the rectifier diodes. Let's imagine that at the initial moment of time the diode VD1 conducts current. The directions of EMF action are shown in the diagram "a" (Fig. 14.9).

Figure: 14.9.Explanation of the effect of the final recovery time of rectifier diodes

When the transistor VT1 is turned on, the EMF changes direction (circuit "b"), the diode VD2 opens. But at the same time, the VD1 diode cannot instantly close. Therefore, the secondary winding turns out to be short-circuited by the diode pair VD1-VD2, which causes current surges in the key element (this is clearly seen in the equivalent circuit of the transformer). The current shape of the primary winding on the combined graph at y \u003d 0.5 will be the same as shown in Fig. 14.10.

Figure: 14.10.The nature of the current of the transformer windings in the case of the presence of ideal and real rectifier diodes

In order to avoid switching surges, it is necessary, firstly, to introduce a pause between the closure of CL1 and the opening of CL2 for a time not less than twice the reverse recovery time of the diode tgr. Secondly, if possible, it is better to abandon conventional diodes and use Schottky diodes.

The voltage across the closed switch transistor is the sum of the supply voltage U nand EMF of the primary half-winding, which is currently open. Since the transformation ratio of these windings is 1 (windings with the same number of turns), the overvoltage on the key transistor reaches 2 U n.Therefore, when choosing a transistor, you should pay attention to the permissible voltage between its power electrodes. It is also necessary to take into account that the current of the key transistor is the sum of the direct load current, recalculated into the primary circuit, and the linearly increasing magnetizing current of the primary inductance. The current is trapezoidal.

When determining the maximum duty cycle in the case of using field effect transistors that switch quickly enough, you need to be guided by the value of the reverse recovery delay of the diodes. The time period during which switching is prohibited:

∆t back= 2t rr.

Fill factor correction:

Maximum fill factor:

When using bipolar transistors and IGBT transistors, the maximum possible duty cycle is reduced due to the off and fall times of these transistors, as well as the characteristic "tail":

Experience shows that 1 fill factor does not exceed 0.45 in the most favorable case.

What else is the difference between a real scheme and an ideal one? The resistances of the open diode and the switch transistor are different from zero. The voltage drop across these elements (and the correction for the transformation ratio) can be taken into account as shown in Fig. 14.11.

a) Rectifier diodes: in the open state, the diode drops on average 0.7 ... 1.0 V (standard diode), or 0.5. ..0.6 V (Schottky diode);

b) Key transistors: if a bipolar transistor or an IGBT transistor is used as a key, the voltage Uke will drop on the key (in saturation mode). Typical saturation voltage is 0.2. 0.5 V. For the MOSFET transistor, you need to calculate the voltage:

A preliminary calculation of the main parameters of the push-pull converter circuit should determine the transformation ratio pand the overall power of the transformer. We have already found out that:

Otherwise (taking into account the voltage drop across the keys and rectifier diodes):

Where - minimum possible supply voltage (set at the beginning of development).

For example, if a battery-powered converter is being designed, this voltage can be taken as the voltage measured at the battery terminals at the end of its life.

It is also necessary to determine the minimum value of the duty cycle γ min, based on the maximum value of the supply voltage (this parameter will be needed when determining the parameters of the smoothing output filter):

Now you can proceed to determining the overall power of the transformer, which is calculated as the half-sum of the power transmitted to the primary winding and received from the secondary windings. In the case of a two-winding transformer, the overall power can be defined as the sum of the load powers and the power consumed for the control circuit (if the converter is built in such a way that the control circuit is powered from the same transformer):

The choice of the required magnetic circuit for the transformer is carried out according to the formula for the overall power, deduced in the section "How the transformer works". Using this formula, we must determine the product SS 0. It should be noted that for push-pull converters it is preferable to use toroidal magnetic circuits, since the transformers wound on them are the most compact. So, the overall power of a transformer wound on a magnetic core of specific dimensions:

where η tr- Transformer efficiency (typical value 0.95 ... 0.97) The developer must fulfill the following condition:

The number of turns of the primary semi-winding can be found by the following formula, which is a form of writing the law of electromagnetic induction:

Secondary half-winding turns:

After that, you need to select the required wire diameter and check the filling of the window with copper. If the coefficient a turns out to be more than 0.5, it is necessary to take a magnetic circuit with a large value of S 0 and recalculate the number of turns.

You can determine the transformer overheating temperature using the following formula:

where ∆ E n - - overheating (T n \u003d T a +∆T n);

T p- transformer surface temperature;

R p- total heat losses (on the active resistance of the winding and in the magnetic circuit);

S cool -outer surface area of \u200b\u200bthe transformer;

α - heat transfer coefficient (α \u003d 1.2 10 -3 W / cm 2 ° С).

After calculating the transformer, you need to make a choice power elements according to the permissible values \u200b\u200bof currents and voltages, to facilitate, if necessary, the thermal regime with the help of heat sinks.

A very important issue that now needs to be considered is the choice of a control circuit for a push-pull pulse source. Not so long ago, all these circuits had to be designed on discrete elements, which gave rise to rather cumbersome and not very reliable solutions. Microassemblies used to control single-ended circuits of stabilizers and converters are not directly suitable for use in push-pull circuits, since you need to have two paraphase outputs controlled by one generator. In addition, the microcircuit must contain a special unit for guaranteed limitation of y in order to prevent emergency situations and through currents. It is desirable to have additional protective shutdown inputs. Recently, a large number of specialized microcircuits have been developed, which already have almost all the necessary nodes.

The TL494 microcircuit (manufactured by Texas Instruments, has a domestic analogue KR1114EU1), which is widely used to control power supplies for computers of the IBM-PC type, is described in detail in the available book. As an example, consider the no less interesting CA1524 microcircuit manufactured by Intersil. This microcircuit contains control and monitoring circuits and functions normally when powered from 8 to 40 V. It can be used as part of any stabilizer and converter circuits described in this book.

The main units of the microcircuit (Fig.14.12):

Temperature compensated 5V reference voltage source;

Precise RC oscillator;

Error amplifier (the difference between the required load voltage and the real voltage at the output of the stabilizer);

Key transistor control circuit comparator;

An error amplifier for a current signal in the primary circuit;

push-pull output stage based on fast bipolar transistors;

Remote on / off control circuit.

Figure: 14.12.Functional units of the CA1524 microcircuit from Intersil

Pulse width control (WID) was discussed by us in the chapter on the chopper stabilizer circuit. In this case, the SHIR scheme works the same way. The only feature is the trigger and the logic circuit, which "route" the control pulses, alternately directing them to one output (transistor Sa), then to the other (transistor Sb). The trigger is synchronized with clock pulses from the master oscillator. Clock pulses have a certain duration, which serves to organize a protective pause between turning off one power transistor and turning on the second. Thus, the duty cycle at max cannot exceed 0.45 (the total pause time for two outputs is 10%). The pause time (dead time) can be adjusted by choosing the appropriate value of the time-setting capacitor Art. The operating frequency of the master oscillator is determined by the ratio of rt and St (the choice of these elements shown in Fig. 14.13 is carried out from the graph, Fig. 14.14). It can be noted that tangible values \u200b\u200bof the pause time are obtained at sufficiently large ratings of the capacity of St. If the timing circuit elements are already selected, the "dead time" can be adjusted within 0.5 ... 5.0 μs by connecting the capacitor Cd to pin 3, as shown in fig. 14.15. The value of this capacitor is in the range of 100 ... 1000 pF. However, the designers of the scheme recommend using this method only as a last resort.

Figure: 14.13.Elements of a frequency-setting circuit Figure: 14.14.Schedule of selection of timing chain elements

Another way to control the dead time is to limit the magnitude of the error amplifier voltage (Fig. 14.16).

The error amplifier (pins 1, 2, 9) has a gain of 80 dB (10000) and can be reduced to the required value by connecting a resistor RL between pins 1 (2) and 9 (depending on whether the direct or inverting circuit is used by the designer of the pulse source). Error amplifier unity gain f- 3 MHz. The designers of the microcircuit note that the error amplifier, not covered by the feedback loop, has a so-called transfer characteristic poleat 250 Hz

(The phase shift between the input and output signal at this frequency reaches 45 degrees). The pole is clearly visible on the graph (Fig. 14.18). This is another reason why an amplifier cannot be used without the feedback circuits shown in Fig. 14.17.

Figure: 14.15.Additional capacitor Q, regulating the "dead time" (a), and the graph for choosing its value (b)

Figure: 14.16.Dead time adjustment method by limiting the magnitude of the error amplifier voltage

Figure: 14.17.Feedback in the error amplifier

An open loop source can turn into a generator. To eliminate the possibility of self-excitation, it is recommended to connect a correction chain to pin 9, as shown in. fig. 14.19.

Figure: 14.18.AFC of the error amplifier Figure: 14.19.Corrective chain eliminating self-excitation

Parameters of the CA1524 microcircuit:

Supply voltage 8 ... 40 V;

The maximum frequency of the master oscillator is 300 kHz;

Instability of the output voltage - no more than 1%;

Temperature instability - no more than 2%;

Capacitance range St - 0.001 ... 0.1 μF;

Resistance range rt - 1.8 ... 120 kOhm;

Error amplifier input offset - 0.5 mV;

Error amplifier input current - 1 μA;

The maximum voltage "collector-emitter" of transistors Sa and Sb -40V;

Current protection is triggered when the current consumption of the microcircuit exceeds 100 mA;

The rise time of the collector current of transistors Sa and Sb -0.2 μs;

The time of the decay of the collector current of the transistors Sa and Sb is 0.1 μs.

The microcircuit also has an external control input (pin 10). Disconnection occurs when a high level is applied (nominal current 0.2 mA).

We will return to the CA1524 microcircuit in the practical development of an experimental push-pull converter, and now we will consider the recently appeared low-power integrated sources built according to a push-pull scheme. The need for a low-power converter appears when it is necessary to obtain a voltage, the source of which has no galvanic connection with the rest of the circuit. For example, digital devices for transmitting information over long lines need such sources. Interference induced in a long line can damage the transmitter and receiver, so the communication line is decoupled using matching transformers or optoelectronic devices. Active line terminators require power.

The second example of using galvanically isolated sources is much closer to the subject of the book. A little later, we will consider the so-called bootstrap control method for push-pull stages. We will see that this circuit needs a source that is galvanically isolated from the common wire. In the dynamic mode, this function, as it turns out, can be successfully performed by a capacitor. But in static mode, you cannot do without a normal source. More recently, this problem was solved with the help of an additional one; windings on a mains transformer, which, of course, did not contribute to a reduction in the size of the circuit. The advent of miniature transducers has gracefully solved this problem.

For example, let's analyze the device of the DCP0115 microcircuit of the firm] Burr-Brown, the functional units of which are shown in fig. 14.20, a appearance - in fig. 14.21. The microcircuit contains a high-frequency generator and a push-pull cascade that operates; with a frequency of 400 kHz. A miniature transformer is connected to the power stage, which, nevertheless, allows 1 W power to be obtained at the load (at an output voltage of 15 V). There is also a soft start circuit and an overtemperature blocking circuit with the ability to recover from a trip. Synchronization pins "(sync in, sync out) are used when the microcircuit works in conjunction with other impulse sourcesavailable in the device. Synchronization avoids frequency beats and reduces radiated RFI. The microsource is made in a DIP-14 package.