High frequency power amplifier microcircuit. High frequency generators

The consumed current is 46 mA. The bias voltage V bjas determines the output power level (gain) of the amplifier.

Fig. 33.11. Internal structure and pinout of TSH690, TSH691 microcircuits

Figure: 33.12. Typical inclusion of microcircuits TSH690, TSH691 as an amplifier in the frequency band 300-7000 MHz

and can be adjusted in the range of 0-5.5 (6.0) V. The transfer coefficient of the TSH690 (TSH691) microcircuit with a bias voltage V bias \u003d 2.7 V and a load resistance of 50 Ohm in the frequency band up to 450 MHz is 23 (43) dB, up to 900 (950) MHz - 17 (23) dB.

Practical inclusion of microcircuits TSH690, TSH691 is shown in Fig. 33.12. Recommended element ratings: C1 \u003d C5 \u003d 100-1000 pF; C2 \u003d C4 \u003d 1000 pF; C3 \u003d 0.01 μF; L1 150 nH; L2 56 nH for frequencies not exceeding 450 MHz and 10 nH for frequencies up to 900 MHz. Resistor R1 can adjust the output power level (can be used for automatic output power control system).

The broadband INA50311 (Figure 33.13), manufactured by Hewlett Packard, is intended for use in mobile communications equipment, as well as in consumer electronic equipment, for example, as an antenna amplifier or radio frequency amplifier. The working range of the amplifier is 50-2500 MHz. Supply voltage - 5 V with current consumption up to 17 mA. Average gain

Figure: 33.13. internal structure of the microcircuit ΙΝΑ50311

10 dB. The maximum power of the signal supplied to the input at a frequency of 900 MHz is no more than 10 mW. Noise figure 3.4 dB.

Typical switching on of the microcircuit ΙΝΑ50311 when powered by a 78LO05 voltage stabilizer is shown in Fig. 33.14.

Figure: 33.14. broadband amplifier on the INA50311 chip

Shustov M.A., Circuitry. 500 devices on analog microcircuits. - SPb .: Nauka i Tekhnika, 2013.-352 p.

Power amplifier 10w

The amplifier is designed to work with a transver with P out up to 1 watt. The exciter load, which ensures stable operation on all ranges, is the resistor R1. The setting consists in setting the quiescent current VT2 within 0.3 A (in the absence of a signal at the input).

A 1 volt signal at the input increases the output power to the antenna to 10 watts. Transmission-reception switching is carried out from an external control circuit, which is closed to the case when switching to transmission. This triggers relay K1 and connects the antenna to the output of the power amplifier. When the control circuit is broken, a positive voltage appears at the base of VT1, which opens it. Accordingly, the VT1 collector is about zero. Transistor VT2 closes. Relay type RPV2 / 7 passport RS4.521.952 Chokes L1 and L2 type D1 (1A) with inductance 30 and 10 μH, respectively. Frame diameter L3 - 15 mm wire PEV2 1.5 mm

Broadband power amplifier

Drozdov VV (RA3AO)

To work in conjunction with an all-band KB transceiver, you can use a broadband power amplifier, circuit diagram which is given in Fig. 1. In the ranges of 1.8-21 MHz, its maximum output power in the telegraph mode at a power supply voltage of +50 V and a load resistance of 50 Ohm is about 90 W, in the range of 28 MHz - about 80 W. The peak output power in the single-sideband signal amplification mode with an intermodulation distortion level of less than -36 dB is about 80 and 70 watts, respectively. With well-chosen transistors of the amplifier, the second harmonic level is less than -36 dB, the third harmonic is less than -30 dB in linear amplification mode and less than -20 dB in maximum power mode.

The amplifier is assembled according to a push-pull circuit on powerful field-effect transistors VT1, VT2. The T1 long line transformer provides a transition from a single-ended excitation source to a balanced input of the push-pull stage. Resistors R3, R4 allow matching the input impedance of the stage with a 50-ohm coaxial line at a VSWR of no more than 1.5 in the range 1.8-30 MHz. Their low impedance gives the amplifier very good self-excitation resistance. To set the initial bias corresponding to the operation of transistors in mode B, the circuit Rl, R2, R5 is used. Diodes VD1, VD2 and VD3, VD4 together with capacitor C7 form a peak detector of the ALC circuit and protection of transistors against overvoltage in the drain circuit. The operation threshold of this circuit is mainly determined by the stabilization voltage of the Zener diode VD9 and is close to 98 V. The VD5-VD8 diodes serve for "instant" protection of the drain circuit from overvoltage. The T3 long line transformer provides a transition from the balanced output of the amplifier to an unbalanced load. To ease the bandwidth requirements of this transformer and reduce possible voltage surges in the drain circuit, a balanced low-pass filter C8L1C10, C9L2C11 with a cutoff frequency of about 30 MHz is connected in front of the transformer.

Mounting the amplifier hinged. The amplifier is assembled on a finned radiator-heat sink made of duralumin with dimensions of 110x90x45 mm. The ribs are milled on both sides of the radiator, their number is 2x13, the thickness of each is 2 mm, the height is 15 mm from the side where the transistors are installed and 20 mm from the side of their fastening nuts. On the longitudinal axis of the radiator, at a distance of 25 mm from the transverse axis, pads with a diameter of 30 mm are milled for installing transistors, and on the reverse side - for fastening nuts. Between the transistors, a "common wire" bus is laid on the radiator fins, cut from 0.5 mm thick sheet copper and attached to the base of the radiator with two M3 screws passed between the two central fins at a distance of 10 mm from its edges. Tire dimensions - 90x40 mm. Mounting posts are attached to the bus. Coils L1 and L2 are frameless and wound with bare copper wire 1.5 mm in diameter on an 8 mm mandrel. With a winding length of 16 mm, they have five turns. Transformer T1 is wound with two stranded wires PEL.SHO 0.31 with a twist pitch of about three twists per centimeter on an annular magnetic core made of M400NN ferrite of standard size K10x6x5 and contains 2x9 turns. Transformers T2 and T3 are wound on circular magnetic cores made of ferrite of the same brand, standard size K32x20x6. Transformer T2 contains 2x5 turns of twisting from wires PELSHO 0.8 with a pitch of two twists per centimeter, T3-2x8 turns of such a twist. Capacitors Cl - C3 - type KM5 or KM6, C4-C7-KM4, C8-C11-KT3.

Establishing a properly assembled amplifier with serviceable parts is reduced to adjusting the inductances of the coils L1 and L2 to the maximum return in the 30 MHz range by compressing or stretching the turns of the coils and to setting the initial offset using resistor R1 to minimize intermodulation distortion in the single-sideband signal amplification mode.

It should be noted that the level of distortion and harmonics largely depends on the accuracy of the selection of transistors. If it is not possible to select transistors with similar parameters, then for each transistor separate circuits for setting the initial bias should be made, and also, at a minimum of harmonics, select one of the resistors R3 or R4 by connecting additional ones in parallel.

In the linear amplification mode in the 14-28 MHz bands, due to the presence of low-pass filters C8L1C10, C9L2C11, the level of harmonics at the amplifier output does not exceed the permissible norm of 50 mW, and it can be connected directly to the antenna. In the 1.8-10 MHz bands, the amplifier should be connected to the antenna through the simplest low-pass filter, similar to the C8L1C10 circuit, and two filters are enough, one for the 1.8 and 3.5 MHz bands, the other for the 7 and 10 MHz bands. The capacitance of both capacitors of the first filter is 2200 pF each, the second one is 820 pF each, the inductance of the first filter is about 1.7 μH, the second is about 0.6 μH. It is convenient to make the coils frameless from bare copper wire with a diameter of 1.5 - 2 mm, wound on a mandrel with a diameter of 20 mm (the diameter of the coils is about 25 mm). The coil of the first filter contains 11 turns with a winding length of 30 mm, the second - six turns with a winding length of 25 mm. The filters are tuned by stretching and squeezing the turns of the coils to the maximum recoil in the 3.5 and 10 MHz bands. If the amplifier is used in overvoltage mode, separate filters should be included on each band.

The amplifier input can also be matched to a 75-ohm coaxial line. For this, the values \u200b\u200bof the resistors R3, R4 are taken at 39 ohms. In this case, the power consumed from the exciter will decrease by 1.3 times, but the gain cutoff at high-frequency ranges may increase. To equalize the frequency response in series with the capacitors C1 and C2, you can turn on coils with an experimentally selected inductance, which should be about 0.1-0.2 μH.

The amplifier can be directly loaded into 75 ohms. Due to the action of the ALC loop, the linear understressed mode of its operation will be preserved, but the output power will decrease by 1.5 times.

Power amplifier on KP904

E. Ivanov (RA3PAO)

When repeating the power amplifier UY5DJ (1), it turned out that the most critical node that reduces the reliability of the entire amplifier is the output stage. After experiments on various types of bipolar transistors, I had to switch to field-effect.

The output stage of the UT5TA wideband amplifier (2) was taken as a basis. The circuit is shown in Fig. 1. the new parts are highlighted with bold lines. A small number of parts made it possible to mount the stage on a printed circuit board and a heatsink from UY5DJ in place of parts and transistors of the UY5DJ amplifier. The quiescent current of the transistors is 100 ... 200 mA.

High-frequency power amplifiers are built according to a circuit containing amplification stages, a filter, and automation circuits. Amplifiers are characterized by nominal output and minimum input power, operating frequency range, efficiency, sensitivity to load changes, level of unwanted vibrations, stability and reliability of operation, weight, dimensions, cost.

The currently obtained maximum output power values \u200b\u200bat frequencies up to 100 MHz are several tens of kilowatts. With a significantly lower power delivered by individual transistors (no more than 200 W), these values \u200b\u200bare achieved by special signal combining devices, among which the most common are power dividers and adders. There are many varieties of these devices. By the magnitude of the phase shift, they are divided into in-phase (with a phase shift of the summed signals φ \u003d 0), antiphase (φ \u003d i), quadrature (φ \u003d n / 2), etc.; by type of execution - with distributed and lumped elements; by the method of connection with the load - for serial and parallel, etc.

One of the main requirements for signal combining devices is to ensure the least mutual influence of individual modules, the powers of which are summed up (the so-called module decoupling). Let's see how this requirement is fulfilled in a simple common-mode transformer adder. The circuit of such an adder on transformers T4- T6together with a divider (on transformers T1- TK)and summed stages (on transistors VT1 and VT2) without bias and power circuits is shown in Fig. 5.4. Transformers T4- T6have transformation ratios, respectively, 1.1 and 1 / V2 (here r n is the load resistance, R B is a ballast resistor, the resistance of which is 2 g n). Under normal operating conditions, when the voltages across the collectors are in phase and their amplitudes are equal, there is no current in the ballast resistor. Transformer T6leads to two series-connected transformer windings T4and T5the resistance is 2r n, so that the load resistance on the collector of each transistor is r n. Imagine now that the collector of the transistor VT2 turned out to be closed with its emitter. In this case, the secondary winding of the transformer T5represents an extremely low resistance for the HF signal, so that the resistance 2r n, reduced to the primary winding of the transformer T6,completely reduced to the secondary winding of the transformer T4, atherefore, to the collector of the transistor VT1. But in parallel VT1 in this case, a ballast resistor of the same resistance is connected, that is, despite the change in the operating mode, in the second stage the operating conditions of the first stage have not changed - it still operates on the load resistance r n. But, since half of its power now goes to the ballast resistor, only half the power of one stage remains in the load, which is 4 times less than the power given by the amplifier to the load before normal operating conditions change. The more stages are used to obtain the output power, the less the change in operating conditions in one stage or another affects the total power in the load. For example, in an amplifier with an output power of 4.5 kW, resulting from the summation of the powers of 32 transistor stages, when one stage fails, the output power decreases to only 4.3 kW. Thus, a very small mutual influence of the stages in the power combining device allows, using the amplifying properties of each transistor to the maximum, to ensure high reliability of its operation, and, consequently, the trouble-free operation of the power amplifier as a whole.

Figure: 5.4. Amplifier circuit with power addition on transformers

The summing device is selected based on the nature and operating conditions of the amplifier, since when solving the main problem - adding signals - it is possible, using certain features of a particular type of adder, to improve other characteristics of the amplifier, for example, to weaken some types of unwanted oscillations or to reduce the sensitivity to load mismatch ...

Satisfactory decoupling of the modules, as well as a low level of unwanted third-order oscillations, low sensitivity to load changes and a weak effect of the summed stages on the preamplifier are obtained when using power quadrature adders. Antiphase combiners suppress unwanted second-order oscillations when decoupled satisfactorily. The alternation of quadrature and antiphase adding devices, for example, when two modules are added in antiphase, and pairs of modules combined in this way are quadrature, largely combines the advantages of both types of adders. For these reasons, quadrature and antiphase adders and power dividers, made, for example, on long coaxial or strip lines, transformers, are widely used in amplifiers with an output power of 10 W and above.

The next parameter of the amplifier - the minimum input power - is determined by the permissible noise level and stability of operation and, in this regard, depends on the circuit, operating mode and design of the amplifier. The effect of noise on amplifier sensitivity is explained as follows. It is known that the noise power brought to the amplifier input is determined by the formula P w \u003d 4kTF w Df, where k - Boltzmann constant; T- absolute temperature; F m - noise figure;

Af is the bandwidth in which the

R sh. But for a given signal-to-noise ratio TO w at the amplifier output power of the input signal R from should not be less than R Sh TO Sh . It follows that the minimum allowable value of the input signal, thus characterizing the sensitivity of the amplifier, is determined as P C tsh \u003d 4kTF u K w Df. Given the TO w and Af all quantities included in this expression are known, with the exception of F JI. With the help of well-known relations, it is easy to show that in a nonlinear amplifier, which in the general case is a power amplifier, with a sufficiently large power gain of the first stage

where F sh1 is the noise figure of the first stage; at t + 1 is the ratio of the noise power amplification factors to the signal power amplification factor in the (m + 1) th stage of the amplifier containing pcascades. Depending on the operating mode of the cascade, this ratio is determined by the formula

coefficients included in this formula are found in tables. For example, for a four-stage 50 W amplifier at F m 1 = 6, Y 2 \u003d 1.6, Yz \u003d 1.7, Y 4 \u003d 1.9 we have F w =31, that at K w \u003d 120 dB, Df \u003d 20 kHz and 4kT \u003d 1.62 * 10-20 W / Hz gives P W \u003d 1 * 10 -14 W and P cmin \u003d 10 MW, i.e., under the specified conditions, the minimum the permissible value of the input signal is characterized by a voltage of about 1 V at a resistance of 75 ohms. Note that the above definition of sensitivity is valid if a signal acts at the input of the amplifier in which the noise power is at least an order of magnitude lower than the amplifier's own noise power reduced to the input, since otherwise an acceptable signal-to-noise ratio will not be obtained Ksh. If this difference in the input noise values \u200b\u200bis not observed, then a selective circuit must be installed between the signal sources and the amplifier to ensure the required K w value, leading to the necessary noise suppression at a given detuning from the operating frequency.

Figure: 5.7. Schemeamplifier with an output power of 15 W for the frequency range 2 - 30 MHz

Table 5.1

Figure: 5.8. Amplifier circuit with 80 W output power for the frequency range 2 - 30 MHz

Table 5.2

The end of the table. 5.2

The bandwidth at high power levels is largely determined by interstage matching circuits, which are specially designed broadband transformers, as well as amplitude-frequency characteristic correction circuits and feedback circuits. So, in fig. Figures 5.7 and 5.8 show amplifier circuits with an output power of 15 and 80 W for radio transmitters with a power of 10 and 50 W operating in the range of 2-30 MHz. Their main characteristics are given in table. 5.1, and the data of the used transformers and chokes - in table. 5.2. The features of these amplifiers are a relatively low level of unwanted oscillations and a relatively small unevenness of the amplitude-frequency characteristic. These parameters, for example, in an 80 W amplifier, are achieved by using frequency-dependent negative feedback in the output stage (from the secondary winding of the transformer TKthrough resistors R11 and R12 to the base of transistors VT3 and VT4) and in the pre-final stage (using resistors R4 - R7), andalso corrective circuits C2 R2, C3 R3 and R1 L1 C1.

It is also possible to reduce the gain flatness in the frequency band by using correction circuits at the input of the final stage (capacitor C7and inductance of conductors ABand VG,which are strips of foil 30 mm long and 4 mm wide) and at the amplifier output (the transformer inductance T4and capacitor C 13). The broadband transformers used in these amplifiers are able to provide satisfactory matching not only in the 2-30 MHz range, but also at higher frequencies. However, at frequencies above 30 MHz, best performance is obtained with strip line transformers without ferrite materials. Such transformers, for example, were used in an amplifier with an output power of 80 W in the range of 30 - 80 MHz (Table 5.3), the circuit of which is shown in Fig. 5.9. A feature of this amplifier is the use of both bipolar and field-effect transistors. This combination made it possible to improve the noise characteristics in relation to the use of only bipolar transistors, and in comparison with the use of only field devices, to improve the energy characteristics of the amplifier.

Table 5.3

Figure: 5.9 Amplifier circuit with 80 W output power for the frequency range 30 --- 80 MHz

An important parameter of an RF amplifier is its efficiency. This parameter depends on the purpose of the amplifier, its operating conditions and, as a consequence, on the construction scheme and the semiconductor devices used. It is 40 - 90% for signal amplifiers with constant or switched amplitude (for example, for frequency and phase modulation, frequency and amplitude telegraphy) and 30 - 60% for linear amplifiers with amplitude modulation. The lower of the indicated values \u200b\u200bare explained by the use of energetically unfavorable, but providing linear amplification, undervoltage modes in all stages, as well as mode A in the preliminary, and often in the pre-final stage of the amplifier. Higher values \u200b\u200bare typical for the key mode of amplification of signals with constant or switched amplitude (80 - 90%) or for amplitude-modulated signals (50 - 60%) when using the method of separate amplification of signal components. For example, an efficiency of at least 80% was obtained in a 4.5 kW broadband amplifier with an output stage on 32 transistors, built taking into account the general recommendations for the key mode and when taking measures to eliminate through currents. However, despite the obvious energy advantages of the key mode of operation, it is still relatively rarely used in RF amplifiers. This is due to a number of features, which, for example, include criticality to load changes, a high level of unwanted oscillations, a high probability of exceeding the maximum permissible transistor voltages and the difficulty of adjusting when obtaining the necessary phase-frequency characteristics, the stability of which must be ensured under conditions of varying load, supply voltage and temperature. environment. In addition, for the implementation of the key mode at high frequencies, transistors with extremely short duration of transients when turning on and off are required.

A promising direction for improving the energy characteristics of amplifiers of an amplitude-modulated signal is signal quantization by level with separate amplification of discrete components and their subsequent summation, taking into account phase shifts.

In increasing the efficiency of amplifiers, the quality of matching with the load, taking into account the possibility of its change, plays an important role. Currently, this issue is simple and at the same time most effectively solved by using ferrite valves and circulators. However, this is the case at relatively high frequencies, at least above 80 MHz. With decreasing frequency, the efficiency of using ferrite decouplers drops sharply. In this regard, it is of interest to study and subsequent industrial development of semiconductor nonreciprocal devices with the properties of circulators, which in principle allow operation at low frequencies. If the use of valves or circulators is not possible, satisfactory results are obtained by combining conventional matching devices with automatic control of the amplifier operation. So, increasing the supply voltage with an increase in the load resistance (with a constant or slightly reduced excitation) and decreasing it with a decrease in the load resistance with an increase in excitation, it is possible to obtain not only a constant output power, but also to maintain the high efficiency value, which was received in nominal mode. The possibilities of this method of stabilizing the output power, however, are limited by the maximum permissible currents and voltages of the used transistor, as well as by the technical capabilities of matching low resistances. For these reasons, the currently implemented region of load resistances, in which a relatively stable output power can still be achieved in this way, is limited, as shown by tests of an amplifier with an output power of 4.5 kW, by a VSWR value not exceeding 3.

The effect of low sensitivity to load mismatch can also be obtained when building an amplifier according to a power addition circuit using quadrature adders and power dividers. With the appropriate excitation voltage, such an amplifier can be achieved, despite the change in the operating mode of each of the summed stages, a slight change in the total consumption current and total output power. When testing such amplifiers, it was noted that the change in output power with a load mismatch turns out to be the same as in linear circuits, i.e., it is described by an expression close to P / P n \u003d 4p / (1 + p) 2, where P n and R- power at nominal and unmatched load, ar - VSWR, characterizing the degree of mismatch. Such a change on average, as shown by comparative tests, is approximately half that of an amplifier built, for example, according to a push-pull circuit.

There are other ways to reduce the sensitivity of the amplifier to load mismatch, but all of them, to one degree or another, are inferior to those considered.

Recently, the level of unwanted oscillations arising in the process of amplifying the useful signal has become one of the main parameters of the amplifier. Such oscillations appear in the power amplifier due to nonlinear processes under the influence of the useful signal f and interference coming from the signal shaping path (f f), the power source (f p) and the antenna of the radio transmitter (f a). Extraneous vibrations (interference) from the signal forming path lead to unwanted emissions of the radio transmitting device not only at the frequencies of these vibrations ff, but also at the frequencies formed under their influence of combination vibrations mf± nf f . The level of such emissions is determined by the relative level of unwanted oscillations at the output of the formation path, its change (transformation) in the power amplifier, as well as the filtering and radiating properties of the radio transmitting device nodes following the amplifier. The change in the noise / signal ratio in the amplifier (K \u200b\u200by) is determined by the switching circuit of the transistor, the operating mode of the stages, the value and frequency of the useful signal and noise.

The largest change in the noise / signal ratio is observed in an amplifier with an OE, as well as at a low output impedance of the signal source r r in an amplifier with OB and at a low load resistance r n in an amplifier with OK. With an increase in r g in an amplifier with OB and r n in an amplifier with O "K K y -\u003e 1. When the amplifier operates in modes A and B with any switching on of the transistor, the relative level of interference does not change; displacement of the operating mode towards mode C leads to an increase, and towards mode AB, on the contrary, to a decrease in the relative level of interference; at the same time, the increase is more noticeable than a decrease.An increase in the mode strength decreases the relative level of interference.The larger the value of the useful signal, the more the interference / signal ratio changes with the same operation mode. the interference / signal ratio decreases.

Combination oscillations arising under the influence of interference are especially dangerous when the amplifier operates in C mode, where their level at the amplifier output is commensurate with the noise level. With a change in the operating mode from C to A, the level of combination oscillations of the second order (f ± ff) decreases monotonically, and the third (2f ± ff) passes through 0 in mode B and upon reaching a minimum in the region of negative values, indicating a change in the phase of oscillations to the opposite , when approaching mode A tends to 0.

All other things being equal, the amplifier with OK is distinguished by the greatest suppression of combination oscillations, and then amplifiers with OB and OE. In a multi-stage amplifier, in contrast to a single-stage amplifier, the interference for each next stage, starting from the second, is not only amplified unwanted oscillations of the formation path, but also combination and harmonic oscillations of the previous stages. The influence of the second harmonic is especially great; it increases the levels of second- and third-order combinational oscillations and decreases the interference / signal ratio. This is mainly manifested in mode C and is actually absent in A. Under its action, the linear operating mode (K y \u003d 1) shifts from mode B to C. These changes are directly opposite if the phase of the second harmonic is somehow artificially changed to l

A low level of combination oscillations, a slight deterioration of the interference / signal ratio and at the same time acceptable energy characteristics are characteristic of an amplifier, the preliminary stages of which operate in modes A - B, and the output stages in modes B - C. When the transistors are turned on according to the OK scheme, modes B - C can be used and in the preliminary stages, but in the output stage, switching on according to the OK scheme is unacceptable due to the high susceptibility of the amplifier to the signals of extraneous radio transmitters. The best for the output stage is the inclusion of the device according to the OB or OE scheme. In this case, the deterioration of the noise / signal ratio in the amplifier at a low level of combination oscillations can be at most 3 dB. But with an illiterate design of the amplifier, this value can increase to 20 dB, and the highest level of unwanted oscillations will be not only at the frequency of the interference, but also at the frequencies caused by this interference of the combination oscillations.

When there is a frequency detuning between the useful signal and the interference, the interference is most effectively suppressed in amplifiers with filters. Suppression is implemented both with electronically commutated filters and by building an amplifier based on a powerful self-oscillator controlled by a phase-locked loop system. In the latter case, it is possible to obtain attenuation of unwanted components - up to 70 - 80 dB, starting from a 5% offset of their frequency from the frequency of the useful signal.

The currently existing transistors in the undervoltage mode of the cascade allow obtaining the level of intermodulation oscillations of the third order - (15 - 30) dB in relation to the interference that caused them when switched on according to the OE scheme, about 15 dB less when switched on according to the OB scheme and, vice versa 15 dB more when switched on according to the OK scheme. Additional suppression of about 15 - 20 dB can be obtained using the quadrature summation of the signals of the modules in the output stage and at least another 15 dB, using a ferrite valve or circulator at the amplifier output.

The highest level of unwanted oscillations is observed at the harmonics of the useful signal. In a single-stage amplifier, without taking any measures to suppress them, this level for the second and third harmonics is usually - (15 - 20) dB. By switching on cascades according to the power addition circuit using quadrature and antiphase adders and dividers, it can be reduced to - (30 - 40) dB. If a filter bank is installed behind the amplifier, this level is reduced by the amount of attenuation of the corresponding filter in the stopband.

Filters can achieve a high level of harmonic suppression. However, it should be emphasized that the harmonics are attenuated; to a level below - 120 dB is possible only with very careful shielding of the RF stages and the elimination of various contact connections in the path after the power amplifier, including RF connectors, in which harmonic oscillations with the same level can form.

As you can see, the existing technical solutions provide high suppression of unwanted vibrations. However, in some cases it still turns out to be insufficient for the normal operation of the equipment. So, when the transceivers located on mobile devices approach each other or when working as part of radio complexes, where a wide variety of equipment is concentrated and must function in extremely limited space, radio receivers often cannot work with their correspondents, as soon as a nearby radio transmitter of another communication line is turned on. This is due to the exposure of the receivers to some unwanted emissions from the radio transmitter. These primarily include noise. Despite the low level, it is they who fly

the greatest danger under these conditions, since, having a continuous spectrum and a weakly varying spectral density with detuning, can, if the necessary measures are not taken, almost completely paralyze the operation of the nearby receivers.

A great danger in this situation is represented by interference from the transmitter signal formation path and the combination oscillations formed by them in the power amplifier, which, like noise, occupy a wide frequency range and cannot be significantly minimized when constructing an amplifier according to the previously considered principle of direct cascade power amplification.

We continue the conversation about the transistor receiver of direct amplification, which we began in the seventh workshop. Connecting then the detector receiver with a single-stage bass amplifier, you thereby turned them into a 0-V-1 receiver. Then I assembled a one-transistor reflex receiver, and at the previous workshop I added a two-stage bass amplifier to it - it turned out to be a 1-V-3 receiver. Now try adding a modulated high frequency (HF) preamp stage to it to become a 2-V-3 receiver. The sensitivity in this case will be sufficient to receive not only local, but also remote broadcasting stations on a magnetic antenna.

What is required for such a single stage RF amplifier? In the main one - a low-power high-frequency transistor of any of the P401 ... P403, P416, P416, P422, GT308 series, if only it is serviceable, several capacitors, a resistor and a 600NN ferrite ring with an outer diameter of 8 ... 10 mm. The coefficient h21E of the transistor can be in the range of 50 ... 100. You should not use a transistor with a large static current transfer coefficient - an experienced amplifier will be prone to self-excitation.

The schematic diagram of the amplifier is shown in Fig. 56. The amplifier itself is formed only by the transistor V1 and resistors R1, R2. Resistor R2 acts as a load, and the base resistor R1 determines the operating mode of the transistor. The collector load of the transistor can be a high frequency choke - the same as in a reflex receiver.

Custom outline L1 C1 and communication coil L2 refer to the input circuit, the capacitor C2 - separating. This part is an exact repetition of the input part of the receiver you have already tested. Capacitor Sraz, resistor R, diode V2, telephones B1 sblocking their capacitor SBL form a detector circuit necessary to test the amplifier.

How does such an amplifier work? Basically the same as a single-stage bass amplifier. He only amplifies the vibrations not of the audio frequency, like that amplifier, but the modulated high-frequency vibrations coming to him from the communication coil L2. The high frequency signal amplified by the transistor is isolated on the load resistor R2 (or other collector load) and can be fed to the input of the second stage for additional amplification or to a detector to convert it to a low frequency signal.

Mount the amplifier parts on a temporary (cardboard) board, as shown on the right in Fig. 56. Transfer here and connect with the amplifier the details of the input circuit (L1C1) and the communication coil (L2) of the receiver. Do not forget to include an isolation capacitor in the coupling coil circuit. C2.Connect the battery voltage 9 inand by picking up the base resistor R1, set the collector current of the transistor within 0.8 ... 1.2 mA. Do not forget: the resistance of the base resistor should be the greater, the greater the static current transfer coefficient of the transistor (the value of this resistor, indicated in the diagram, corresponds to the coefficient h21E transistor about 50).

Now, on a separate small cardboard box, mount the detector circuit by connecting in series the phones B1 with the blocking capacitor Cbl with a capacity of 2200..3300 pF, a point diode V2 any series and separator capacitor Сraz with a capacity of 3300 ... 6800 pF, Resistor resistance R maybe 4.7 ... 6.8 kOhm. Connect this circuit between the collector and the emitter of the transistor, that is, to the output of the amplifier, and connect an outdoor or indoor antenna and, of course, ground to the input circuit of L1C1. When tuning the input circuit to a local radio station, its high-frequency signal will be amplified by a transistor VI, diode detected V2 and converted by phones IN 1into the sound. Resistor R in this circuit is necessary for the normal operation of the detector. Without it, phones will sound quieter and with distorted sound.

On the day of the next experiment with an RF amplifier, a high-frequency step-down transformer is needed (Fig. 57). Wind it on a 600NN ferrite ring (the same as the high-frequency choke core of the receiver's reflex stage). Its primary winding L3 should contain 180 ... 200 turns of wire PEV or PEL 0.1 ... 0.12, and the secondary L 4 60 ... 80 turns of the same wire.

Turn on the L3 winding of the high-frequency transformer in the collector circuit of the transistor instead of the load resistor, and to its winding L4 connect the same detector circuit as in the previous experiment, but without the blocking capacitor and resistor, which are not needed now. How does it sound now? phones? Louder. This is explained by the better, than in the first experiment, matching the output impedance of the amplifier and the input impedance of the detector target.

Now, using the diagram shown in Fig. 58, connect this single-stage amplifier to the transistor input of the 1-V-C reflex receiver. The RF receiver amplifier has become a two-stage amplifier. The coil became the connecting element between the cascades L4 high-frequency transformer included in the base circuit of the transistor V 2 (in the receiver 1-V-З it was a transistor W1) instead of a communication coil (there was L2) with the former input tunable circuit. Now an external antenna and grounding are not needed - the reception is carried out on the W1 magnetic antenna. the role of which: performs a ferrite core with a coil on it L1 input tunable loop L1 C1.

So, together with a two-stage LF amplifier, I learned a four-transistor direct amplification receiver 2-U-Z. The receiver may be self-excited. This is because, firstly, it is reflexive, and reflex receivers are generally prone to self-excitation, and secondly, the conductors connecting the experimental amplifying cascade with the reflexive cascade are long. If the new stage together with the magnetic antenna is mounted compactly on the same receiver board, making the circuits as short as possible, there will be fewer reasons for self-excitation. The decoupling filter cell also contributes to this. R2 C3 in the negative supply circuit of the first transistor of the RF amplifier, which eliminates the connection between the stages through a common source of litany and thereby prevents self-excitation of the high-frequency channel of the receiver.

But the second stage of the RF amplifier can be the same as the first, that is, not reflex, and the connection between them may not be transformer. The diagram of a possible amplifier is shown in Fig. 59. Here the load of the transistor V1 the first stage, as in the first experiment of this workshop (see Fig. 56), is the resistor R2; The voltage of the high-frequency signal created on it through the capacitor SZfed to the base of the transistor V2 the second stage, exactly the same as the first. The signal, additionally amplified by the second stage transistor, is removed from its load resistor R4 (the same; like R 2) and through capacitor C 4 (such as SZ)goes to the detector on the diode V 3, is detected by it, and low-frequency oscillations created on its pull-up resistor R5, are fed to the input of the bass amplifier.

In this version, the second stage and the detector represent, as it were, an unfolded reflex stage of the previous version. But the transistor only amplifies high frequency oscillations. And if you connect it to a two-stage bass amplifier, you get a direct amplification receiver 2- V-2. The amplification of the low-frequency signal will decrease slightly, the phones or the loudspeaker head at the output of such a receiver will sound a little quieter, but the danger of self-excitation of its high-frequency path will decrease. This loss can be partially compensated by an increase in the voltage of the low-frequency signal at the output of the detector by including a second diode in the detector stage (in Fig. 59 - shown by dashed lines V4), as you did in one of the experiments of the seventh workshop (see Fig. 50), or use a transistor in the detector stage.

Try to experiment with the options for the bass amplifier, compare the quality of their work to draw appropriate conclusions for the future.

One more tip. Experimenting with this or that version of the receiver, devils and memorize its complete schematic diagram. What for? A radio amateur, even a beginner, must draw diagrams of such devices from memory. The schematic diagram, in addition, will help you better understand the operation of the receiver as a whole and its parts, and will facilitate troubleshooting in it.

Literature: Borisov V.G. Practical work of a beginner radio amateur. 2nd ed., Revised. and add. - M .: DOSAAF, 1984.144 p., Ill. 55k.

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